The present invention relates in general to temperature-compensated electronic reference circuits and components therefor, and is particularly directed to a new and improved voltage-controlled current generator, which is operative to generate an output current that exhibits a prescribed non-linear to linear characteristic with temperature when its control voltage range is restricted. Injecting this output current into a voltage reference circuit, such as a xe2x80x98Brokawxe2x80x99 bandgap voltage reference, provides improved high-order curvature correction, yielding an output voltage whose variation over a temperature range (e.g., xe2x88x9220xc2x0 C. to +125xc2x0 C.) is extremely flat (e.g., within several hundreds of microvolts).
FIG. 1 is a reduced complexity diagram of a conventional first-order, current-based bandgap voltage reference, which generates an output voltage that is substantially independent of temperature, by summing a plurality of components whose temperature coefficients vary in a mutually complementary manner. For this purpose, a current I1 proportional to absolute temperature (PTAT) is supplied to a series circuit of a diode D1 and a resistor R1 (referenced to ground (GND)). The voltage V1 across diode D1 has an inverse or complementary to absolute temperature (CTAT) characteristic. As a result, if the (PTAT) current I1 is inversely proportional to the value of a resistor having the same temperature coefficient as that of the resistor R1, the temperature behaviors of the respective voltage drops across diode D1 and resistor R1 will be mutually complementary, making the output voltage V2 at a terminal OUT substantially (to a first order) independent of temperature.
A non-limiting example of what is commonly referred to as a xe2x80x98Brokaw cellxe2x80x99 current mirror implementation of the temperature-compensated bandgap voltage reference of FIG. 1 is schematically shown in FIG. 2. A current mirror circuit is formed of a first pair of MOSFETs including a MOSFET M1 and a diode-connected MOSFET M2 in a current mirror first leg containing a diode-connected MOSFET M5, and a second pair of MOSFETs comprised of diode-connected MOSFET M3 and MOSFET M4 in a second current mirror leg containing a MOSFET M6. MOSFETs M1 and M4 have their source-drain paths coupled in series with those MOSFETs M5 and M6 between voltage rail VDD and GND.
Diode-connected MOSFET M2 has its gate connected in common with the gate of MOSFET M1, while MOSFET M4 has its gate connected in common with the gate of diode-connected MOSFET M3. MOSFET M2 has its source-drain path coupled in series with the collector-emitter path of a bipolar NPN transistor Q1 and resistors R2 and R3 to GND. In a complementary manner, MOSFET M3 has its source-drain path coupled in series with the collector-emitter path of a bipolar NPN transistor Q2 and resistor R3 to GND. The bases of transistors Q1 and Q2 are coupled to a voltage output terminal OUT. A MOSFET M7 has its source-drain path coupled between voltage rail VDD and output node OUT, to which an output resistor RL referenced to GND is coupled. MOSFET M7 has its gate coupled to the drain of MOSFET M4.
In the current mirror-based implementation of FIG. 2, the current flowing through MOSFETs M2 and M3 corresponds to the base-emitter difference voltage xcex94VBE divided by the value of resistor R2, and is PTAT. Thus, the current I1 supplied through resistor R3 produces a PTAT voltage thereacross which is combined with the CTAT VBE voltage V1 across transistor Q2 to derive an output voltage reference V2 having a first-order compensated temperature coefficient. As shown in FIG. 3, the voltage V2 of the bandgap reference circuit of FIG. 2 varies with temperature in a substantially parabolic manner, and has a total variation on the order of 6.7 mV. A first order voltage references of the type shown in FIG. 2 is capable of producing a reference voltage whose temperature coefficient typically falls between 20 to 100 ppm/xc2x0 c.
FIG. 4 illustrates a high-order compensating modification of the current-based voltage reference of FIG. 1, which employs an additional current component I2 having a non-linear temperature coefficient. This additional non-linear current is intended to compensate for high-order, temperature dependent terms from the contribution of voltage V1. In the voltage reference of FIG. 4, the resistor R1 of reference circuit of FIG. 1 is shown as series-connected resistors R4 and R5, with an additional, high-order compensation non-linear current I2 being supplied to the common connection of these two resistors.
One example of this type of bandgap voltage reference circuit that injects an additional non-linear current is disclosed in the U.S. Pat. No. 6,157,245 to Rincòn-Mora. (For a non-limiting example of additional prior art documentation showing another type of current-based bandgap reference circuit, attention may be directed to the U.S. Pat. No. 5,952,873 to Rincòn-Mora.) The above-referenced ""245 patent describes that second-order compensation may be achieved by injecting an additional non-linear current whose temperature coefficient is proportional to the xe2x80x98squarexe2x80x99 of PTAT (or I2PTAT), so that its characteristic is parabolic.
While the squared temperature coefficient of the additional current tends to provide a second-order improvement at relatively low temperatures, where the variation in slope of the parabolic (squared) temperature coefficient characteristic is relatively gradual, the performance of an I2PTAT current-based voltage reference undesirably degrades at higher temperatures, due to the increasingly steep slope of the injected current""s parabolic characteristic at such temperatures.
In accordance with the present invention, shortcomings of conventional first- and second-order compensated voltage references, including those described above, are substantially reduced by injecting into a voltage reference circuit, such as a xe2x80x98Brokawxe2x80x99 voltage reference, a high-order, compensation current derived from a voltage controlled, non-linear current generator. This non-linear current generator is configured to generate an output current whose temperature coefficient exhibits a prescribed non-linear-to-quasi-linear curvature when the input or control voltage range is restricted. As will be described, this particular current characteristic enables a voltage reference that incorporates such a non-linear current generator for high-order curvature correction to produce an output voltage whose variation over an operational temperature range (e.g., xe2x88x9220xc2x0 C. to +125xc2x0 C.) is extremely flat (e.g., within several hundreds of microvolts). The inclusion of the non-linear current generator described herein in a bandgap reference allows a simple, power and area efficient method to achieve a curvature corrected output voltage.
To this end, the non-linear current generator according to the invention comprises an input transistor, referenced to a first power supply rail and having its collector-emitter path coupled in series with a PN junction device, such as a diode-connected transistor, to series-connected resistors that are coupled to a second power supply rail. The control electrode or base of the input transistor is coupled to receive an input or xe2x80x98referencexe2x80x99 (control) voltage, whose value is restricted or maintained within an xe2x80x98optimumxe2x80x99 range, in accordance with the desired operational parameters of the diode-connected transistor. In particular, this control voltage is set to a value, such that, in the low temperature region of operational temperature range, the diode-connected bipolar transistor operates just below the non-linear transition or xe2x80x98kneexe2x80x99 of its non-linear transfer characteristic. An output transistor has its emitter coupled to the common connection of the series resistors and its base coupled in common with the base of the diode-connected transistor. The collector of the output transistor is coupled to a current mirror, which mirrors the non-linear collector current from the output transistor as the desired non-linear output current INL.
At cold temperatures and with an input voltage such that the voltage drop across the base-emitter of the input and output transistors causes the resistance of each branch to be large, the output current is very small. With an increase in temperature, the characteristics of the bipolar junction transistor cause the resistance of each collector-emitter path to decrease in an exponential fashion. As a consequence, the voltage across a summation resistor increases in the same exponential fashion and so does the output current. As temperature increases, the resistance of the collector-emitter paths of the transistors of the two branches becomes comparable to the resistance of the summation resistor, allowing some of the voltage drop from the input voltage to ground to be applied across the summation resistor.
The resistance of the series resistor is set such that it becomes larger than the decreasing collector-emitter resistance of the diode-connected transistor, so that its branch resistance stops its exponential decrease and becomes dependent on the resistance of the resistor in series with the summation resistor. The effect of the resistance of the diode-connected transistor and the series resistor branch being dominated by the series resistor, and thus the output transistor resistance becoming comparatively smaller, is such that the base-emitter voltage of the output transistor begins to decrease with temperature.
With the decrease in the base-emitter voltage of the output transistor, its collector-emitter path resistance begins to increase again, until the effects of increasing temperature become more dominant again and cause the resistance to decrease. At temperatures above this point, the voltage across the summation resistor increases in proportion to the temperature coefficient of 2VBEs, which is on the order of (xe2x88x921)(2)(xe2x88x922 mV/xc2x0 C.)=xc2x14 mV/xc2x0 C., on a first-order basis.
The characteristics of the output current of the non-linear current generator improve the temperature performance of the bandgap voltage reference. With a first-order bandgap voltage reference curve shifted toward colder temperatures, the added positive temperature coefficient of the non-linear current generator initially causes the decreasing output voltage to increase. Then, as the slope of the output current vs. temperature of the non-linear current generator begins to decrease, the output voltage starts to decrease, until the contribution of the non-linear current causes the output voltage to increase again. When the resistor values are properly chosen, an optimized output voltage temperature characteristic can be realized.
Summing the voltages across the series resistors and the base-emitter junction of the bipolar transistor whose base and emitter are coupled between the voltage output terminal and the series resistors causes the curvature-corrected reference voltage circuit, to which the non-linear current INL is injected, to produce a reference voltage whose variation with temperature is confined within a very narrow (several hundred microvolts) range, which corresponds to a relatively small temperature coefficient (on the order of 2 ppm/xc2x0 C.).
In addition to being used as a source of high-order compensation current for a xe2x80x98Brokawxe2x80x99 current mirror-based bandgap voltage reference, the non-linear current generator of the invention may be combined with other temperature-controlled current sources, to produce a high-order, temperature compensated output current reference IREF, which exhibits an output current vs. temperature variation, that is extremely narrow (e.g., on the order of only tens of nanoamps over a range of from xe2x88x9220xc2x0 C. to 125xc2x0 C.)